Concurrent dual-band receiver architecture

ABSTRACT

The present invention discloses an architecture for a concurrent dual-band high-frequency receiver. The invention combines a concurrent dual-band front-end subsystem having a dual-band antenna, dual-band pre-amplifier filter and concurrent dual-band LNA with a novel image rejection downconverter to provide the functions of a typical receiver, including reception, amplification and downconversion of a signal in two discrete desired frequency bands simultaneously.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims the benefit of U.S. Provisional Application No.60/275,894, filed Mar. 14, 2001.

FIELD OF THE INVENTION

This invention relates to the field of high frequency receivers and moreparticularly to architectures for wireless receivers that can operate attwo frequency bands simultaneously.

BACKGROUND OF THE INVENTION

Wireless communications systems have exhibited remarkable growth overthe past decade. Wireless voice and data applications are being enabledby rapidly emerging wireless technologies, such as cellular telephony,personal communications systems, bluetooth and wireless local areanetworks (WLAN's), to name a few. Digital modulation techniques,miniaturization of transceivers due to advances in monolithic integratedcircuit designs and the development of high frequency, microwave andmillimeter wave RF systems in both the licensed and unlicensed bands,have all contributed to improving the quality and bandwidth capacity ofthese systems and to reducing the size and costs of the components.

These systems are having a profound effect on societies. For example,they are enabling service-based economy work forces to become“untethered” from their information sources and conventional wiredcommunications mechanisms. Moreover, throughout the world, wirelesscommunication systems are enabling developing countries to provideinstant telephone service to new subscribers who otherwise would have towait years for wireline access.

Dual band receivers have been introduced to the marketplace thatincrease the functionality of such communication systems. Thesereceivers can receive only one band at a time and thus must switchbetween the two different bands. FIG. 1 is a conceptual schematic ofsuch a conventional non-concurent, heterodyne dual band architecture 10.As seen, an incoming signal, V_(in), is received at a switch 12 (forsimplicity the antenna and front-end filter are not shown). If thesignal is in a first predetermined frequency band, ω₁, the switch movesto the top signal processing path tuned to match and amplify signalsonly in this band. The signal is then impedance matched and amplified atlow noise amplifier (“LNA”) 20, filtered at band pass filter (“BPF”) 21,mixed with local oscillator signal, LO₁, at mixer 22, filtered at BPF₂24 and mixed again with a second local oscillator signal, LO₂, at mixer26, until it exits as V_(out1) (e.g. baseband or some low frequency) forfurther processing (e.g. digital signal processing). If the incomingsignal is in the second predetermined frequency band, ω₂, the switch 12moves to the bottom signal processing path tuned to match and amplifysignals only in this band. In particular, the signal is amplified by LNA30, filtered at BPF₃ 31, mixed with a third local osillator tuned toω′_(LO1) at mixer 32, filtered again at BPF₄ 34 and mixed again with afourth local oscillator tuned to ω′_(LO2) at mixer 36 and exists asV_(out2). In the example shown, the four oscillators are completelyindependent devices. While such functionality adds to a device'sversatility, such as in the case of a dual-band digital cellular phone,these receivers are very inefficient in terms of component parts andpower consumption and would not satisfy the needs for thenext-generation of multi-functional devices, such as a cell phone with aGPS receiver and a bluetooth interface.

Another problem with conventional wireless technology relates tobandwidth limitations. The diverse range of modern wireless applicationsdemand wireless communications systems and transceivers with greaterbandwidth capacity and flexibility than can be conventionally supplied.Increased bandwidth capacity is necessary for many wireless applicationsto become a reality. Wireless broadband Internet applications (e.g.browsing, e-commerce, streaming audio and video), wireless videomessaging, wireless video games, and remote video monitoring are just afew examples of applications that will be delivered over the nextgenerations of wireless networks. Conventional solid-state radiofrequency (“RF” or “wireless”) receiver architectures, such assuperheterodyne and direct conversion receivers, accomplish highselectivity and sensitivity by designing them for narrow-band operationat a single RF frequency. Unfortunately, these modes of operation are oflimited functionality because they limit the system's availablebandwidth and robustness to channel variations. On the other hand,wide-band modes of operation are more sensitive to out-of-band signalsdue to transistor non-linearity, which can introduce severe bottlenecksin system performance.

Thus, to overcome these and other drawbacks, it would be highlydesirable to have a low cost, concurrent dual-band receiver. As usedherein a concurrent dual-band receiver is one that can process signalsat two discrete frequency bands simultaneously, or substantiallysimultaneously. This would enable a receiver to significantly increaseits bandwidth capacity (bit rate). A concurrent dual-band receiverdesign could also be used for supplying redundancy in mission criticaldata transmission application. The reliability of the received signalwould be greatly increased with simultaneous transmission of the samesignal in multiple bands, because channel properties are different anduncorrelated at two frequency bands and more diversity is achieved.

A further challenge for modem receiver design is to create trueconcurrent dual-band functionality using as little real estate (andideally monolithically) and as little power dissipation as possible (andperhaps no more than single band receivers), while keeping theincremental production costs above the conventional single band receiverto a minimum.

SUMMARY OF THE INVENTION

The present invention, which addresses these needs, resides in aconcurrent, dual-band receiver architecture that is capable ofsimultaneous operation at two different frequencies without dissipatingtwice as much power or a significant increase in cost and footprint.This concurrent operation can be used to extend the available bandwidth,provide new functionality and/or add diversity to battle channel fading.These new concurrent multi-band receivers provide at multiple frequencybands simultaneous 1) narrow-band gain and matching, and 2) imagerejection downconversion.

In accordance with the present invention a concurrent, dual-bandreceiver that receives a signal at two discrete desired frequency bands,simultaneously, is disclosed. The receiver includes a concurrent,dual-band, front-end subsystem and a concurrent, dual-band,image-rejection, frequency downconverter. The front-end subsystemproduces an RF signal having signal attenuation regions at frequencybands outside the two desired frequency bands.

In particular, the downconverter includes first and subsequentimage-rejection downconversion stages. The first downconversion stagereceives the RF signal from the front end subsystem and is adapted tosimultaneously downconvert the RF signal at two frequency bands to twointermediate frequency (IF) bands such that the image frequency bands ofthe two desired frequency bands fall at the attenuation region of thefront-end transfer function. The subsequent image rejectiondownconversion stage downconverts the two IF bands down to eitherbaseband or to a desired low frequency.

More particularly, the front-end subsystem includes a concurrent,dual-band antenna, a concurrent, dual-band bandpass filter connected tothe antenna that receives the dual-band signal from the antenna, and aconcurrent, dual-band LNA connected to the filter that providessimultaneous gain and impedance matching at the multiple bands whilemaintaining a relatively low noise figure.

Also, disclosed is a concurrent, dual-band, image-rejectiondownconverter for a concurrent dual-band RF receiver having a front-endsubsystem that supplies a front-end signal having two discrete desiredfrequency bands and attentuation regions at frequency bands outside thetwo discrete desired frequency bands. The downconverter includes a firstimage-rejection downconversion stage that receives and that is adaptedto simultaneously downconvert the front-end signal to two intermediatefrequency (IF) bands such that the image frequency bands of the twodesired frequency bands fall at the attenuation regions of the front-endsignal, and a subsequent image-rejection downconversion stage thatdownconverts the two IF bands.

In one preferred embodiment, the first downconversion stage includes afirst quadrature local oscillator (LO₁) block and two mixers. The LO₁block is adapted to supply an in-phase (I) signal and a quadrature (Q)signal of a first predetermined frequency. The first mixer is connectedto the LO₁ block and the front-end subsystem and is adapted to mix thein-phase LO₁ signal with the front-end signal and to supply a resultantin-phase intermediate frequency (IF) signal. The second mixer isconnected to the LO₁ block and the front-end subsystem and is adapted tomix the quadrature LO₁ signal with the front-end signal and to supply aresultant quadrature IF signal. The first predetermined frequency of theLO₁ block is offset from the midpoint of the two desired bands such thatthe image frequency bands of the two desired bands fall at attenuationregions of the front-end signal. In this way, the images aresignificantly attenuated prior to the subsequent amplification,filtering and downconversion stages.

In the preferred embodiment, the downconverter further includes anin-phase IF filtering and amplification stage connected to the firstmixer and a quadrature IF filtering and amplification stage connected tothe second mixer.

In a more detailed embodiment of the present invention, the subsequentdownconversion stage is a second and last stage. It includes a secondquadrature local oscillator (LO₂) block adapted produce an in-phase (I)signal and a quadrature (Q) signal at a second given frequency, a firstintermediate frequency (IF) mixing stage connected to the LO₂ block andthe first mixer that is adapted to mix the in-phase intermediatefrequency (IF) signal with the in-phase signal of the LO₂ block and tosupply a resultant first low frequency (LF) signal, and a second IFmixing stage connected to the LO₂ block and the second mixer that isadapted to mix the quadrature signal of the LO₂ block with thequadrature IF signal and to supply a resultant second LF signal. It alsoincludes a third quadrature local oscillator (LO₃) block adapted producean in-phase (I) signal and a quadrature (Q) signal at a third givenfrequency, a third IF mixing stage connected to the LO₃ block and thefirst mixer that is adapted to mix the in-phase LO₃ signal with thein-phase IF signal and to supply a resultant third LF signal, and afourth IF mixing stage connected to the LO₃ block and the second mixerthat is adapted to mix the quadrature LO₃ signal with the quadrature IFsignal and to supply a resultant fourth LF signal.

The downconverter further includes a first summing circuit that combinesthe first and second LF signals to constructively add the first desiredbaseband signal and destructively combine the baseband image signalassociated with the first desired baseband signal. The downconverteralso includes a second summing circuit that combines the third andfourth LF signals to constructively add the second desired basebandsignal and destructively combine the baseband image signal associatedwith the second desired baseband signal. In yet more detail, the firstIF mixing stage includes a first IF mixer, the second IF mixing stageincludes a second IF mixer, the third IF mixing stage includes a thirdIF mixer and the fourth IF mixing stage includes a fourth IF mixer.

However, in an even more detailed embodiment of the present invention,the downconverter's the first IF mixing stage includes a first IF mixerthat mixes the in-phase IF signal with in-phase LO₂ signal, and a fifthIF mixer that mixes the in-phase IF signal with quadrature LO₂ signal,the second IF mixing stage includes a second mixer that mixes thequadrature IF signal with the quadrature LO₂ signal and a sixth IF mixerthat mixes the quadrature IF signal with the in-phase LO₂ signal, thethird IF mixing stage includes a third mixer that mixes the in-phase IFsignal with the in-phase LO₃ signal and a seventh mixer that mixes thein-phase IF signal with the quadrature LO₃ signal, and the fourth IFmixing stage includes a fourth mixer that mixes the quadrature IF signalwith the quadrature LO₃ signal and an eighth mixer that mixes thequadrature IF signal with the in-phase LO₃ signal.

In this more detailed embodiment, the downconverter further includes afirst summing circuit that sums the outputs of the first and secondmixers, a second summer that sums the outputs of the fifth and sixthmixers, a third summing circuit that sums the outputs of the third andfourth mixers, and a fourth summing circuit that sums the outputs of theseventh and eighth mixers.

In a more detailed aspect of the first downconversion stage of thepresent invention, a front-end signal phase shifter is connected to thefront-end subsystem that provides a quadrature front-end signal along aquadrature front-end signal path. The stage also includes a first localoscillator (LO₁) block dapted to supply an in-phase (I) signal of afirst predetermined frequency, a first mixer connected to the LO₁ blockand the front-end subsystem that is adapted to mix the LO₁ signal withthe front-end signal and to supply a resultant in-phase intermediatefrequency (IF) signal; and a second mixer connected to the LO₁ block andthe front-end phase-shifter and adapted to mix the LO₁ signal with thequadrature front-end signal and to supply a resultant quadrature IFsignal.

A method of concurrently dowconvertering a dual-band RF signal is alsodislcosed. The method includes providing a dual-band RF signal having afront-end transfer function having two desired frequency bands andattenuation notches at undesired frequency bands, and downconverting theRF signal such that the image frequencies of the two-bands fall at theattenuation notches of the front end transfer function.

More particularly, the downconverting step includes splitting the RFsignal to first and second signal processing paths, mixing the RF signalon the first path with an in-phase first local oscillator (LO₁) signalto produce an in-phase intermediate frequency (IF) signal, filtering thein-phase IF signal, mixing the RF signal on the second path with aquadrature LO₁ signal to produce a quadrature IF signal, filtering thequadrature IF signal, mixing the filtered in-phase IF signal with anin-phase second local oscillator (LO₂) signal, mixing the filteredquadrature IF signal with the a quadrature LO₂ signal, mixing thefiltered in-phase IF signal with an in-phase LO₃ signal; mixing thefiltered quadrature IF signal with the quadrature LO₃ signal; adding themixed in-phase LO₂ signal to the quadrature LO₂ signal, and subtractingthe mixed in-phase LO₃ signal from the mixed quadrature LO₃ signal.

Other features and advantages of the present invention should becomemore apparent from the following description of the preferredembodiments, taken in conjunction with the accompanying drawings, whichillustrate, by way of example, the principles of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a conceptual schematic of a conventional, non-concurrentdual-band receiver architecture that is designed to switch from one bandto the other;

FIG. 2 is conceptual schematic of a concurrent dual-band receiver thathas two independent paths;

FIG. 3 is a schematic of a non-concurrent dual band receiver that usesthe “Weaver” image-rejection architecture;

FIG. 4 is a conceptual schematic showing a primary goal of the presentinvention, namely, a concurrent dual-band front-end subsystem of areceiver feeding a concurrent dual-band image rejection downconverterhaving two ouputs;

FIG. 5 is a schematic of one proposed implementation of a concurrentdual band receiver, using the “Weaver” image-rejection technique for thedownconverter stage;

FIG. 6 is schematic showing one preferred embodiment of the concurrentdual-band receiver of the present invention, wherein the downconvertershown in FIG. 4 is a modified “Weaver” image-rejection circuit forconcurrent dual-band downconversion;

FIG. 7(a) is a frequency domain graph showing frequency response of theconcurrent, dual-band front-end with signals at the point just prior tothe first downconversion stage in FIG. 6;

FIG. 7(b) is a graph of the frequency domain signal of the concurrent,dual-band receiver of FIG. 6 after the first downconversion stage, butbefore the second downconversion stage;

FIG. 7(c) is a graph of the frequency domain signal of the dual bandreceiver of FIG. 6 at the output of the downconverter;

FIG. 8 is a schematic showing a specific implementation of a concurrentdual band 2.45 GHz/5.80 GHz receiver designed according to the presentinvention;

FIGS. 9(a)-(c) are graphs of the transfer functions of the receiverbuilding blocks and frequency domain representation of the signalsreceived by the receiver shown in FIG. 8;

FIG. 10 is a general model for a single transistor amplifier witharbitrary impedances connected to the terminals;

FIG. 11 is an equivalent small-signal model of the transistor amplifiershown in FIG. 10, with the transistor shown as a combination ofimpedance and voltage dependent current sources;

FIG. 12 is a simplified schematic of FIG. 11 with the active device(transistor) disposed on an ac-grounded bulk substrate;

FIG. 13 is an input impedance network system schematic of the presentinvention showing the input impedances of the amplifier looking into thegate of the transistor;

FIG. 14 is a schematic of a concurrent dual band CMOS LNA;

FIG. 15 is a graph showing the measured voltage gain and S₁₁ of thedual-band LNA shown in FIG. 14;

FIG. 16 is an illustration of the crossband intermodulation of a dualband LNA; and

FIG. 17 is a micrograph of a concurrent dual band LNA.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The invention summarized above and defined by the enumerated claims maybe better understood by referring to the following detailed description,which should be read in conjunction with the accompanying drawings. Thisdetailed description of particular preferred embodiments, set out belowto enable one to build and use particular implementations of theinvention, is not intended to limit the enumerated claims, but to serveas a particular examples thereof. The particular example set out belowis the preferred specific implementations of a concurrent dual-band highfrequency RF receiver and methods for designing the same. It should beunderstood however, that this system and technique is not limited todual-band receivers, but may be extended to other applications, such astransmitters and transceivers. It should also be understood that theterm “concurrent” refers to the ability to process a signal having twofrequency bands of interest simultaneously or substantiallysimultaneously.

Using conventional receiver technology, a way to provide concurrentdual-band functionality is to design into a receiver two independentsignal paths with two sets of components (antennas, bandpass filters,LNA's, downconverters, etc.), each tuned for a discrete frequency band.For example, a dual-band receiver 40 using two parallel heterodynereceiver paths is shown schematically in FIG. 2. As shown, this designis similar to the dual band receiver architecture shown in FIG. 1without the switch 12. Moreover, as as a further improvement, as isknown in the art, the first and second local oscillators, LO₁ and LO₂,are shared by the top and bottom signal paths affording a savings of twolocal oscillators as compared with the design in FIG. 1. The frequencyof LO₁, ω_(LO1), is $\frac{\omega_{1} + \omega_{2}}{2}$and the frequency of LO₂, ω_(LO2), is$\frac{\omega_{1} - \omega_{2}}{2}.$This scheme is essentially equivalent to designing multiple single band,heterodyne receivers, each tuned to a different band and stuffed intoone package. Unfortunately, this architecture significantly increasesthe cost, footprint and power dissipation of a receiver, as comparedwith a single path architecture.

A primary objective of the present invention is to create, as shown inFIG. 4, a concurrent dual-band receiver 100 that comprises a concurrentdual-band front-end subsystem 110 connected to a concurrent, dual banddownconverter 120. It should be understood that, preferably, in order toachieve a truly concurrent dual-band receiver with maximum efficiencyusing minimum real estate and power consumption, as many components ofthe receiver as possible, namely, the antenna 112, pre-LNA filter 114,and LNA 116 of the front end subsystem, as well as the downconverter120, should function as concurrent dual-band components. The receivershould have two separate outputs V_(out1) and V_(out2), that cansubstantially simultaneously provide baseband or near baseband outputsextracted from the two desired frequency bands. The front-end subsystemand downconverter of such a receiver will now be discussed.

1. The Dual-band Front End Subsystem

A concurrent, dual-band front-end subsystem became possible with theadvent of the concurrent multi-band LNA, disclosed in pending U.S.patent application Ser. No. 09/821,403, filed on Mar. 14, 2001 andtitled “Concurrent Multi-Band Low Noise Amplifier Architecture” by theinventors of the present invention. A very important observation madethere was that the transconductance of the active device of the LNA iswideband and can be used to provide gain and matching at otherfrequencies of interest without any penalty in power dissipation. Thisobservation led to several topologies for dual-band LNA's that providesimultaneous gain and matching at two bands (as seen in the graphs ofFIG. 4), triple band LNA's and more generally multi-band LNA's. Adetailed description of the dual-band LNA is described in sections 4 and5, below.

The other components of a concurrent dual-band front-end subsystem havebeen disclosed in the art. For example, concurrent dual-band antennaswere disclosed in D. M. Pozar et al., “A Dual Band Circularly PolarizedAperture-Coupled Stacked Microstrip Antenna for Global PositioningSatellite,” IEEE Trans. Antennas Propagation, vol. 45, no. 10, pp.1618-1625, October 1997; Zi D. Liu et al., “Dual-Frequency PlanarInverted-F Antenna,” IEEE Trans. Antennas Propagation, vol. 45, no. 10pp. 1451-1458, October 1997; and L. Zaid, et al., “Dual-Frequency andBroad-Band Antennas with Stacked Quarter Wavelength Elements,” IEEETrans. Antennas Propagation, vol. 47, no. 4, April 1999. Moreover, amonolithic dual-band filter has also been disclosed in H. Miyake et al.,“A Miniaturized Monolithic Dual Band Filter Using Ceramic LaminationsTechnique for Dual Mode Portable Telephones,” IEEE MicrowaveMillimeter-Wave Monolithic Circuits Symp. Dig., pp. 789-792, 1997. Thus,the components for the entire front-end subsystem of the concurrentdual-band receiver are available.

2. The Dual-band Downconverter

The present invention completes the missing piece of the concurrentdual-band receiver by disclosing a novel, concurrent, dual-band,image-rejection downconverter. In the preferred embodiment, thisdownconverter receives the amplified and filtered dual-band signal fromthe front-end subsystem and converts each band substantiallysimultaneously first to an intermediate frequency and then to a lowenough frequency to be processed while rejecting substantially all ofthe image frequencies.

Numerous downconversion schemes are known in the art. One such family ofschemes is known as single sideband image rejection. One well-knownimage rejection scheme disclosed by Weaver (See Weaver, Jr., D., “AThird Method of Generation and Detection of Single Sideband Signals,”Proceedings of the IRE, pp. 1703-1705, June 1956), and hereinafterreferred to as “the Weaver architecture”, is now reviewed. The Weaverarchitecture performs two consecutive quadrature downconversionoperations on the received signal and its image is fed from thefront-end such that if the final outputs are added together the signalis obtained and the image is suppressed, and if the outputs aresubtracted, the image is obtained and the signal is suppressed. Thissolution was applied by Wu, et al. in “A 900-MHz/1.8 GHz CMOS Receiverfor Dual-Band Applications,” IEEE Journal of Solid-State Circuits, Vol.3, No. 12, (December 1998), for the design of a non-concurrent dual-bandreceiver 50. As shown in FIG. 3, in this design, the frequency of thefirst local oscillator LO₁ 60 is chosen halfway between the twofrequencies bands and the particular frequency of interest isalternately selected by choosing the appropriate sideband produced by animage separation mixer.

As stated in Wu, et al., the signal received by the antenna 52 in eachband is applied to a duplexer filter 54, 56 to perform band selection.Subsequently, a low noise amplifier (LNA) 62, 64 and two quadraturemixers 66, 68 and 70, 72 boost and translate the signal to an IF of 450MHz. The results of the two bands are combined at this IF and undergo asecond quadrature downconversion operation as in the conventional Weaverdesign. The LNA and RF mixers of the two bands are separate to allowflexibility in the choice of device dimensions and bias currents, thusoptimizing the performance of each path independently. For the seconddownconversion, two sets of quadrature downconversion mixers 84, 86 havebeen used to provide both I and Q baseband outputs. The bandpass filters80, 82 are formed by means of on-chip inductors and parasiticcapacitances, but they do not perform channel selection.

In the receiver of FIG. 3, the first LO frequency is set midway betweenthe GSM and DCS1800 bands, making the two bands images of each other.That is, the RF mixing uses high-side injection for GSM and low-sideinjection for DCS1800. The band-select input switches the receiverbetween the two operating modes (GSM or DCS 1800), shutting off the RFpath of the idle band to save power consumption. Also, the band-selectswitch 90 controls the addition or subtraction at the receiver output inorder to generate the desired signal and reject the image component.

While the Weaver architecture by itself does not provide sufficientimage rejection, the 900-MHz spacing between the signal and the imageallows substantial image filtering in the front-end duplexers.

Applying the Weaver architecture in the way Wu and Razavi did for aconcurrent dual-band receiver, however, is not a satisfactory solution.This will be best understood with reference to the exemplaryconfiguration shown in FIG. 5. In particular, a dual-band antenna 202,selectively and simultaneously receives a signal in two frequency bands.The dual-band signal is filtered by a dual-band bandpass filter (BPF)204 and simultaneously amplified and impedance matched at bothfrequencies by a concurrent dual-band LNA 206. A quadraturedownconverter stage splits the received signal down two parallel paths212 and 214, respectively. In path 212, the received signal is mixedwith a reference signal from a first local oscillator (LO₁) 208 at mixer220 to produce the in-phase (I) component of the received signal.Following Wu and Razavi, the frequency of LO₁, or ω_(LO1), is set halfway between the center frequencies of the two desired bands of thesystem. Along path 214, the received signal is mixed with the referenceat mixer 222, wherein the reference signal from LO₁ 208 is phase shiftedby ninety degrees at phase shifter 210, producing the quadrature (Q)component of the received signal. The in-phase component (I) and thequadrature component (Q) are then further amplified and filtered atblocks 224 and 226, respectively, and are mixed in a second stage atmixers 234 and 236, respectively, with an in-phase LO₂ signal fed fromoscillator block 230 and a quadrature LO₂ signal fed from the oscillatorblock, respectively. The two paths are then combined in a way to produceconstructive interference on the signal of interest to produce signal Aand destructive interference on the image signal.

In the concurrent downconversion scheme, however, since the unwantedimage signal is one of the two desired signal bands, there is noattenuation of the image by any of the antenna, the front-end bandpassfilter or the dual-band LNA. Thus, one must rely solely on the imagerejection of Weaver's single sideband downconverter, which is limited bythe phase and amplitude mismatch of the quadrature local oscillators andsignal paths, and can only provide about 20-40 dB attenuation of theunwanted image in each band. This is clearly insufficient imagerejection for the intermediate frequency signals and thus fails as asolution to the concurrent dual-band problem.

A solution is derived from analyzing the dual-band front-end subsystem'stransfer function (at point “a” in FIGS. 4 and 6) shown in FIG. 7(a). Asseen, the dual-band transfer function just prior to downconversion hastwo clearly distinct amplification (or desired frequency) bands, hereincalled bands “A” and “B”, separated by a signal attenuation region.There are also attenuation regions below band A and above band B, for atotal of three attenuation regions. These attenuation regions arecompletely determined by the compounded attenuation of the dual-bandantenna, filter and LNA.

By offsetting the first local oscillator frequency LO₁ from the midpointbetween bands A and B, as shown in the figure, applying the Weaver imagerejection technique now not only does not suffer from the aforementioneddrawbacks, but actually significantly improves the image rejection. Thekey to this solution is to offset the LO₁ frequency of the first stageof the image-rejection architecture from the midpoint of the two bandsof interest in such a way that the image, f_(IA), of the first band,f_(A), falls at the middle attentuation region of the front-endsubsystem transfer function. Similarly, the image of the second, upperdesired band, f_(B), falls at outside the pass-band of the front-end atf_(IB) and will also be attenuated.

The inventive concurrent dual-band receiver architecture that implementsthis technique is shown in FIG. 6. The receiver 300 has a front-endsubsystem having a dual-band antenna, dual band bandpass filter 304 anddual-band LNA 306 to provide the dual-band front end signal at “a”having the transfer function shown in FIG. 7(a). This signal is splitinto two paths. A first quadrature local oscillator block 310 having apredetermined frequency f_(LO1) that satisfies this offset frequencyrequirement feeds a pair of mixers on the separate paths, just like thefirst half of any single-sideband image reject architecture, such as theWeaver architecture, and results in a pair of downconverted intermediatefrequency (IF) signals, IF_(A) and IF_(B), as shown in FIG. 7(b), withrelatively small image components superimposed on the two signals.

At least one subsequent downconversion stage is needed to further rejectthe remaining image signal and to take the signals down to baseband ornear baseband. In the preferred embodiment shown in FIG. 6, only oneadditional stage is needed. Thus, IF_(A) traveling on the upper path 308and IF_(B) traveling on the lower path 309 are passed through low passfilter/amplifier blocks 316 and 318, respectively, to eliminate theunwanted upconverted signals and to amplify the desired downconvertedsignals.

The two signals, herein called the in-phase intermediate frequency (IF)signal, IF₁, and quadrature intermediate frequency signal, IF_(Q), areeach split into two separate paths and are concurrently fed into thesubsequent downconversion stage to produce baseband, or near baseband,signals, A and B. In the preferred embodiment shown in FIG. 6, a secondquadrature local oscillator, LO₂, 320 having a frequency f_(LO2) andthird quadrature local oscillator, LO₃, 340 having a frequency f_(LO3)are provided. Along the upper path, the in-phase LO₂ signal is mixedwith IF_(I) at mixer 322 and the quadrature LO₂ signal is mixed withIF_(Q) at the second mixer 324. The resultant signals are then combinedat summing circuit 330 to produce the first desired baseband signal, A,shown as the first graph of FIG. 7(c).

Along the lower path, the in-phase LO₃ signal is mixed with IF_(I) atmixer 342 and the quadrature LO₃ signal is mixed with IF_(Q) at a fourthmixer 344. These two resultant signals are then recombined at summingcircuit 350 to produce the second desired baseband signal, B, shown asthe second graph of FIG. 7(c).

It will be understood by those skilled in the art that the finaldownconversion signal need not be at baseband. The signal couldalternatively be downconverted to any appropriate frequency (i.e. nearbaseband) that is capable of being further processed by subsequentstages.

Note that in practical implementations, three separate frequencysynthesizers to generate the three mentioned local oscillators are notneeded. Since two channels (in two bands) are independently selected,only two frequency synthesizers are needed. A third local oscillator canbe generated from (one of) those. For example, the second localoscillator 440 shown in FIG. 8 is generated by dividing the outpoutfrequency of synthesizer 450 by 4.

It should be understood that variations to the circuit shown in FIG. 6are within the scope of the present invention. For example, asillustrated in the specific concurrent dual-band receiver design circuitshown in FIG. 8, each mixer in the second downconversion stage mayalternatively comprise a mixing stage that includes more than one mixer.

3. A Concurrent 2.45 GHz/5.80 GHz Receiver

The architecture of the present invention has been implemented in thedesign of a concurrent dual-band receiver that simultaneously receivesand downconverts signals in the 2.45 GHz band and 5.80 GHz band. Thecircuit block diagram for this design is shown in FIG. 8 and itsconceptual transfer functions at the front-end, intermediate frequenciesand outputs with frequency planning details are shown in the graphs ofFIGS. 9(a)-9(c). In particular, the receiver 400 includes a dual-bandfront-end subsystem 410 having a front-end transfer function shown inFIG. 9(a), and a dual-band, single-sideband image rejectiondownconverter 420 designed according to the present invention. Thefront-end subsystem includes a dual-band antenna 412, dual-band filter414 and dual band LNA 416, each tuned to pass signals in the 2.45 GHz(f_(A)) and 5.80 GHz (f_(B)) bands.

The front-end signal is split into an upper and lower path. The upperpath is mixed at mixer 434 with an in-phase local oscillator signalproduced by a first local oscillator circuit (frequendy synthesizerloop) 430 having a local oscillator LO₁ 431 and an RC polyphase circuit432. This circuit produces an in-phase intermediate frequency signal,called IF_(I). The lower path is mixed at mixer 436 with the quadratureLO₁ signal produced by LO₁ 431 and polyphase circuit 432 and produces aquadrature intermediate frequency signal, called IF_(Q). As seen in FIG.9(a), the frequency of the first local oscillator LO₁ 430 isstrategically set at 3.10 GHz, so that when mixed with receiveddual-band signal, the image of f_(A), or f_(IA), will be at 3.75 GHz,which as seen in FIG. 9(a) is advantageously at an attentuation regionof the front end transfer function. Similarly, the image of the secondband, f_(IB), is at 0.40 GHz, which is also at an attentuation region ofthe front-end transfer function.

IF_(I) and IF_(Q) are then further filtered and amplified at 438 and439, respectively, with an IF transfer function as shown in FIG. 9(b).The downconverter now processes the two paths in the second, and in thispreferred embodiment, final, downconversion stage as follows.

A second quadrature local oscillator (LO₂) block 440 is to produce anin-phase (I) signal (f_(LO2I)) and a quadrature (Q) signal (f_(LO2Q)) ata second given frequency. In the present embodiment, as shown in FIG.9(b), this frequency is 0.65 GHz, the same as the frequency of IF_(A)signal. A third quadrature local oscillator (LO₃) block is also providedto produce an in-phase (I) signal (f_(LO3I)) and a quadrature (Q) signal(f_(LO3Q)) at a third given frequency. This frequency is set at 2.70GHz. Since in the preferred embodiment, f_(LO2) is one fourth f_(LO3),block 440 is shown as a “divide-by-four” circuit, hence there will be noneed for more than two frequency synthesizers.

Following the architecture described in conjunction with FIG. 6, thesecond downconversion stage also includes four intermediate frequency(IF) mixing stages. In particular, a first IF mixing stage 460 mixes thefiltered and amplified upper path IF_(I) signal with the in-phasesignal, f_(LO2I), of the LO₂ block at a first mixer 462 and supplies aresultant first low frequency (LF₁) signal. A second IF mixing stage 470mixes f_(LO2Q) with the lower path IF_(Q) signal at mixer 472 andsupplies a resultant second low frequency signal (LF₂). LF₁ and LF₂ aresummed at summing circuit 495 to produce the first in-phase basebandsignal, denoted as BB,I_(A).

A third IF mixing stage 480 mixes f_(LO3I) with the IF_(I) at a thirdmixer 482 and supplies a resultant third low frequency signal LF₃, and afourth IF mixing stage 490 mixes f_(LO3Q) with the lower path IF_(Q)signal and to supply a resultant fourth low frequency signal, LF₄. LF₃and LF₄ are summed at summing circuit 497 to produce the second desiredin-phase baseband signal, denoted as BB,I_(B). In the preferedembodiment, these signals are at baseband, but need not be.

However, in the preferred embodiment shown in FIG. 8, each of the fourIF mixing stages includes an additional mixer to mix the variouscombinations of in-phase and quadarature signals. By properly combining(add/subtract) these mixed signals, not only are quadrature signals atbaseband provided, but improved image rejection is also achieved. Asdisclosed by Crols and Steyaert in “A Single-Chip 900 MHz CMOS ReceiverFront-End With A High Performance Low-IF Topology,” IEEE Journal ofSolid-State Circuits, Vol. 30, No. 12, December 1995, this technique isknown in the art as double quadrature downconversion. In particular,mixing stage 460 includes a fifth mixer 464 that mixes f.sub.LO2Q withIF.sub.I and mixing stage 470 includes a sixth mixer 474 that mixesf.sub.LO2I with IF.sub.Q. The outputs of these mixers 464 and 474 arecombined at summing circuit 496 to produce a first quadrature basebandsignal, denoted as BB,Q.sub.A. Moreover, mixing stage 480 includes aseventh mixer 484 that mixes f.sub.LO3Q with IF.sub.1 and mixing stage490 includes an eighth mixer 494 that mixes f.sub.LO3I with IF.sub.Q.The outputs of these mixers, 484 and 494 are combined at summing circuit498 to produce a second quadrature baseband signal, denoted asBB,Q.sub.B.

4. Description of a Concurrent Dual-band Low Noise Amplifier

Traditional single-band LNA's use a single or cascode transistor stageto provide wide-band transconductance and combine it with proper passiveresonant circuitry at the input and output to shape the frequencyresponse and achieve gain and matching at the single band of interest.See e.g., Shaffer et al., “A 1.5-V, 1.5 GHz CMOS Low Noise Amplifier,”IEEE JSSC, vol. 32, No. 5, pp. 745-59, May 1977. The inventors haveobserved that the wide-band transconductance of the active device can beused to provide gain and matching at other frequencies of interestwithout any penalty in power dissipation. This observation has led theinventors to the concurrent dual-band LNA of the present invention thatprovides simultaneous gain and matching at two bands.

The following provides a generic approach to the design of a generalclass of integrated, single path concurrent multi-band LNA's as one ofthe essential building blocks of concurrent multi-band receivers. In asingle-band LNA, passive circuits are used to shape the widebandtransconductance of the active device in the frequency domain to achievegain and matching at the frequency of interest. This concept can begeneralized to multiple frequency bands noting that the intrinsictransconductance of the active device is inherently wideband and can beused at multiple frequencies simultaneously.

FIG. 10 shows the general case impedance model of a three terminalactive device having an input terminal, an output terminal and a currentsource terminal. The active device shown here is an NFET transistorhaving a gate 1020, g, as its input terminal, a drain 1040, d, as itsoutput terminal, and a source 1060, s, as its current source terminal.The transistor and impedance terminology and symbology used hereinafterfollow the FET transistor convention. However, it should be understoodthat this general case and the specific examples set forth hereinafterapply equally to other types of three terminal active devices, such asbipolar, MESFET, PHEMT transistors, etc.

This general model shows an LNA input signal, V_(in), with an arbitraryseries impedance between the incoming input signal and the gate, Z_(g),a gate-source impedance, Z_(gs), a source impedance Z_(s), a gate-drainimpedance Z_(gd), (also known as the feedback impedance Z_(f)) and aload impedance Z_(L). The impedances shown in FIG. 10 also includetransistor's inherent reactance components (e.g., C_(gs)). This model isredrawn in FIG. 11 with the transistor shown as a combination of currentsources and a drain-to-source resistance, r₀, and disposed on siliconsubstrate bulk, b, with the added impedances introduced from eachtransistor terminal to the bulk, namely, Z_(gb), Z_(bs), and Z_(bd).FIG. 12 is the same as FIG. 11 but with the bulk set to AC ground. Asnoticed in FIG. 12, the bulk-to-source impedance, Z_(bs), can becombined with the source impedance Z_(s) to result in Z′_(s). Further,the bulk-to-drain impedance, Z_(bd), can be combined with the externallyadded load impedance Z_(L) resulting in Z′_(L).

The three primary design considerations for a concurrent dual-band LNAof the present invention are (1) input impedance matching; (2) noisefactor minimization; and (3) output gain. Each of these are nowconsidered in detail.

A. Multiband Input Impedance Matching

An important feature of an LNA is its input impedance matching formaximum power transfer. Neglecting r₀ shown in FIG. 12, the inputadmittance (inverse of impedance) looking into the gate of thetransistor, has been derived by the inventors and is given by theequation: $\begin{matrix}{Y_{i\quad n} = {\frac{1}{Z_{g} + Z_{gs} + {Z_{s}^{\prime}\left( {1 + {g_{m}Z_{gs}}} \right)}} + \frac{1}{Z_{L}^{\prime} + Z_{f}} + \frac{g_{mb}}{1 + \frac{Z_{f}}{Z_{L}^{\prime}}} + {\frac{1}{1 + \frac{Z_{f}}{Z_{L}^{\prime}}} \times \left( {g_{m} - g_{mb}} \right) \times \frac{Z_{gs}}{Z_{gs} + {Z_{s}^{\prime}\left( {1 + {g_{m}Z_{gs}}} \right)}}}}} & {{Eq}.\quad(1)}\end{matrix}$

For purposes of impedance matching, the inventors have designed thebroad schematic shown in FIG. 13 showing the input impedance of the LNAlooking into the gate of the transistor. Converting FIG. 13 to anequation, the input impedance can be defined as the sum of five,two-terminal, frequency-dependent, impedance networks:Z _(in) =Z ₁ +Z ₂ +Z ₃ +Z ₄ +Z ₅  Eq. (2)

wherein

Z₁=Z_(g)+Z_(gs)+Z′_(s)+g_(m)Z′_(s)Z_(gs);

Z₂=Z′_(L)+Z_(f′);

Z₃=[1+Z_(f)/Z′_(L)]/g_(mb), wherein g_(mb) is the bulk effecttransconductance;${Z_{4} = {\frac{1}{g_{m} - g_{mb}} \cdot \left( \frac{Z_{f}}{Z_{L}^{\prime}} \right) \cdot \frac{Z_{gs} + {Z_{s}^{\prime}\left( {1 + {g_{m}Z_{gs}}} \right)}}{Z_{gs}}}};{and}$

Z₅ is the intrinsic gate-to-bulk impedance, Z_(gb).

Further, neglecting the effect of the gate-to-drain impedance Z_(gd)(Z_(f)), that is to say, assuming the transistor's internal reactanceC_(gd) approximates 0 (compared to the other impedances in the network),as well as Z_(gb), the input impedance of the amplifier shown in FIGS.4-8 is simplified toZ _(in) =Z ₁ =Z _(g) +Z _(gs) +Z′ _(s)(1+g _(m) Z _(gs))  Eq. (3)

To achieve a maximum input power match at more than one frequency bandsimultaneously, the inventors have determined that Equation 3 shouldsatisfy the following equation set:Z _(g) +Z _(gs) +Z′ _(s)=0  Eq. (4)

and thus,

 g _(m) Z′ _(s) Z _(gs) =Z _(in) =R _(in)=50Ω,  Eq. (5)

wherein R_(in)=50Ω is the predetermined characteristic impedance of theantenna. It should be understood that any other antenna design having adifferent impedance value could be used. However, as is well understood,50Ω has become a de facto standard in antenna receiver design.

To demonstrate the validity of these expressions, consider the specialcase of a single band LNA inductive source degeneration similar to thatof Shaffer et al., discussed above, where (4) reduces to:$\quad\left\{ \begin{matrix}{{\left( {L_{g} + L_{s}} \right)C_{gs}\varpi^{2}} = 1} \\{\frac{g_{m}L_{s}}{C_{gs}} = {R_{i\quad n} = {50\Omega}}}\end{matrix} \right.$

in accordance with Shaffer et al.

The general design criteria given by Eq. (4) can be used to generate alarge number of different topologies for concurrent multi-band LNAs. Thesection titled “Examples” below presents just two examples of suchtopologies, one for a dual-band LNA and another for a triple-band LNA.

B. Noise Factor

Ignoring the noise contribution of passive elements, the total noise ofan LNA can be represented by its input equivalent voltage and currentnoise: $\begin{matrix}{{i_{n} = {\frac{i_{nd}}{g_{m}Z_{gs}} + i_{ng}}}{e_{n} = {{\frac{Z_{gs} + Z_{s}^{\prime} + Z_{g}}{g_{m}Z_{gs}} \cdot i_{nd}} + {\left( {Z_{s}^{\prime} + Z_{g}} \right) \cdot i_{ng}}}}} & {{Eq}.\quad(6)}\end{matrix}$

where i_(nd) and i_(ng) are the drain and gate noise currents (collectorand base noise currents in a bipolar implementation), and g_(m) is thetransconductance of the transistor.

To obtain more insight into the design trade-offs, the inventors of thepresent invention ignored the gate noise (that usually contributes lessthan 0.2 dB to the NF), in the expression for the noise factor, F, thatis given by: $\begin{matrix}{F = {{1 + \frac{\overset{\_}{{{i_{n} + {Y_{s}e_{n}}}}^{2}}}{\overset{\_}{i_{s}^{2}}}}\quad \approx {1 + {\frac{\gamma\quad g_{d0}}{Y_{s}} \cdot \frac{1}{g_{m}^{2}{Z_{gs}}^{2}} \cdot {{1 + {Y_{s}\left( {Z_{gs} + Z_{s} + Z_{g}} \right)}}}^{2}}}}} & {{Eq}.\quad(7)}\end{matrix}$

where g_(d0) is the zero-bias drain-source channel conductance, Y_(s) isthe reference source admittance (e.g., Y_(s)={fraction (1/50)}Ω) for thenoise figure, NF, i_(s) is the noise current associated with this sourceadmittance, and γ is the excess noise factor for the MOS transistorranging from ⅔ for long-channel devices to more than 2 for short-channeldevices.

Several useful design implications can be obtained from Eq. (7). Firstof all, this equation agrees with the well-accepted notion that NF canbe reduced using a larger g_(m) (more power dissipation). It also showsthat an increase in Z_(gs) improves the NF, that accounts for theimprovement in noise figure for transistors with smaller channel lengthand C_(gs). The last term in Eq. (7) plays the most important role inthe design of concurrent multi-band LNA's. Since passive componentscannot produce any negative real part, the last term reaches its minimumwhen Z_(g)+Z_(gs)+Z′_(s)=0 at the frequency(ies) of interest. Thus, theminimum NF will be achieved for these frequency(ies).

It is thus observed that in order to achieve both minimal noise andmaximum power match at the input for multiple frequencies, equations (3)and (7) should simultaneously satisfy the minimum NF and input matchingcondition at all frequencies of interest. Interestingly, equations (4)and (5) do just that. In addition to these conditions, it is crucial tomaximize Z_(gs) and g_(m) to minimize NF as much as the power budgetallows.

C. Narrow Band Output Gain

In order to achieve narrow-band gain at the bands of interest, the drainload network should exhibit high impedance only at those frequencies ofinterest. Using the model of FIGS. 10-13, the overall gate to drainvoltage gain, Av, (neglecting the body effect and r₀) is given by theequation: $\begin{matrix}{A_{V} = {\frac{Z_{L}Z_{f}}{Z_{L} + Z_{f}} \cdot \left\lbrack {\frac{1}{Z_{f}} - \frac{g_{m}Z_{gs}}{Z_{gs} + {Z_{s}^{\prime}\left( {1 + {g_{m}Z_{gs}}} \right)}}} \right\rbrack}} & {{Eq}.\quad(8)}\end{matrix}$

Again neglecting the feedback impedance Z_(f), (i.e. Z_(f)≈∞) we obtain:$\begin{matrix}{A_{V} = {Z_{L}^{\prime} \cdot \left\lbrack {- \frac{g_{m}Z_{gs}}{Z_{gs} + Z_{s}^{\prime} + {g_{m}Z_{s}^{\prime}Z_{gs}}}} \right\rbrack}} & {{Eq}.\quad(9)}\end{matrix}$

Applying the parameters of Eq. (4), the voltage gain equation simplifieseven further toA _(v) =−Z′ _(L) /Z′ _(s)  Eq. (10)

As discussed below, several resonant circuits satisfy this equation formaximum voltage gain.

5. A Concurrent Dual-band CMOS LNA Topology Example

A large number of passive networks satisfy the design criteria ofEquations 4, 5 and 10. In order to minimize the NF, one should maximizeZ_(gs), as previously mentioned. One way to obtain a reasonably largeZ_(gs), is to use a transistor with minimum channel length and no extrapassive element between the gate and the source. Equation 5 can besatisfied using a single on-chip source degenerative inductor. FIG. 14shows a concurrent dual-band CMOS LNA designed according to the criteriaset forth above (with biasing not shown). In other to fulfill Equation 4at both center frequencies, as shown, a parallel LC network in serieswith the inevitable inductance of the bonding wire and package lead isused. The parallel LC of Z_(g) resonates with Z_(gs)+Z_(s) at bothfrequency bands of interest such that Equation 4 is satisfied.

In order to achieve narrow-band gain at bands of interest, the drainload network should exhibit high impedance only at those frequencies.This can be done by adding a series LC branch in parallel with theparallel LC tank of a single-band LNA, as shown off the drain of thecascode transistor in FIG. 14. Each series LC branch introduces a zeroin the gain transfer function of the LNA at its series resonantfrequency.

It should be understood that this is but one topology that satisfies theabove-derived equations for the design of a concurrent multi-band LNA.Many other topologies can be used.

(i) Concurrent Dual-band LNA Measurement Results

A concurrent dual-band CMOS LNA operating at 2.45 GHz and 5.25 GHzfrequency bands for indoor wireless communications was designed based onthe topology of FIG. 14. This section presents the measurement results.It was implemented in a 0.35 μm BiCMOS technology using only CMOStransistors. The input parallel resonator uses Cg=0.9 pF porcelainmultilayer capacitor and Lg=2.7 nH chip inductor. The inductance of thebonding wire, Lbond=3 nH and the source inductor, Ls=0.7 nH. Turning tothe load network, the high impedance at each of the two frequencies isobtained by providing the series LC branch, C1=240 fF and L1=9.8 nH andthe parallel LC tank, L2=2.3 nH and the inherent parasitic capacitanceof the transistor, which is equivalent of to approximately 300 F.

FIG. 15 shows the measured voltage gain, A_(v), and input reflectioncoefficient, S₁₁, of the amplifier up to 10 GHz. It achieves narrow-bandvoltage gains of 14 dB and 15.5 dB, input return losses of 25 dB and 15dB, and noise figures of 2.3 dB and 4.5 dB at 2.45 GHz and 5.25 GHz,respectively. It drains 4 mA of current from a 2.5V supply voltage. Thenotch due to the LNA is about 40 dB deeper than the peaks which directlytranslated to the same amount of improvement in image rejection. Due tothe large difference between the notch and pass-band frequencies, noelaborate tracking loops such as those proposed by Samavati et al. in “A5-GHz CMOS Wireless LAN Receiver Front End” IEEE JSSC, vol. 35, no. 5,pp. 765-72, May, 2000, are necessary. The single-ended nature of the LNAmakes external Baluns unnecessary. Measurements of 6 different chipswith 3 different boards and off-chip components showed strongrepeatability without using the commonly-used sliding capacitor inputmatching adjustment.

An LNA's linearity is often measured by intermodulation and compressionpoint tests and represented by IP3, for 3rd order non-linearity, and CP1for 1 dB compression point. We refer to these in-band IP3 and CP1, asIP3 _(inband) and CP1 _(inband). However, in a multi-band system, morenon-linearity measures should be considered. In-band signals fromdifferent desired bands (e.g., 2.50 GHz and 5.15 GHz) can mix due toamplifier's non-linearity, causing in-band undesired signals (e.g.,3×2.50−1×5.15=2.35 due to 4th order non-linearity), as shown in FIG. 16.The inventors showed this cross-band IPn, as IPn_(crossband), where n isthe order of non-linearity. A similar cross-band compression measure canbe defined as the signal power in band A that causes a 1 dB drop in thesmall signal gain in band B and vice versa, which will be denoted as CP1_(A>B).

This concurrent dual-band LNA demonstrates an input-referred in-band IP3of 0 dBm and 5.6 dBm, and in-band CP1 of −8.5 dBm and −1.5 dBm at 2.45GHz and 5.25 GHz bands, respectively. The measured input referred IP4_(crossband) is 7.5 dBm. The LNA exhibits an CP1 _(2.4>5.2) of −11.5 dBmand an CP1 _(5.2>2.4) of −5.7 dBm.

The following table summarizes the measured performance of thefabricated monolithic concurrent dual-band LNA shown in FIG. 17. Thechip occupies an area of 0.8×0.8 mm² including pads and ESDs. The NF,S₁₁ and power dissipation are better than previously publishednon-concurrent and/or single-band CMOS LNAs.

Frequency 2.45 GHz ± 50 MHz 5.25 GHz ± 100 MHz Voltage Gain 14 dB 15.5dB S₁₁ −25 dB −15 dB NF 2.3 dB 4.5 dB Input IP3_(in-band) 0.0 dBm 5.6dBm Input CP1_(in-band) −8.5 dBm −1.5 dBm Input CP1_(A>B) CP1_(2.4>5.2)= −11.5 dBm CP1_(5.2>2.4) = −5.7 dBm Input IP4_(cross band) 7.5 dBm DCCurrent 4 mA Supply Voltage 2.5 V Active Device 0.35 μm CMOS transistors

The present invention provide a generic approach to the design of ageneral class of concurrent dual-band receivers. Having thus describedexemplary embodiments of the invention, it will be apparent that furtheralterations, modifications, and improvements will also occur to thoseskilled in the art. Further, it will be apparent that the presenttechnique and system is not limited to any particular pair of frequencybands of interest or specific implementation.

1. A concurrent, dual-band, image-rejection downconverter for aconcurrent dual-band RF receiver having a front-end subsystem thatsupplies a front-end signal having two discrete desired frequency bandsand a transfer function with attentuation regions at frequency bandsoutside the two discrete desired frequency bands, comprising: (a) afirst image-rejection downconversion stage that receives and that isadapted to simultaneously downconvert the front-end signal to twointermediate frequency (IF) bands such that the image frequency bands ofthe two desired frequency bands fall at the attenuation regions of thefront-end transfer function, and further comprising: (i) a firstquadrature local oscillator (LO₁) block adapted to supply an in-phase(I) signal and a quadrature (Q signal of a first predeterminedfrequency; (ii) a first mixer connected to the LO₁ block and thefront-end subsystem that is adapted to mix the in-phase LO₁ signal withthe front-end signal and to supply a resultant in-phase intermediatefrequency (IF) signal; and (iii) a second mixer connected to the LO₁block and the front-end subsystem and adapted to mix the quadrature LO₁signal with the front-end signal and to supply a resultant quadrature IFsignal, wherein the first predetermined frequency of the LO₁ block isoffset from the midpoint of the two desired bands such that the imagefrequency bands of the two desired bands fail at attenuation regions ofthe front-end transfer function; and (b) a subsequent image-rejectiondownconversion stage that downconverts the two IF bands.
 2. Thedownconverter of claim 1, further including an in-phase IF filtering andamplification stage connected to the first mixer and a quadrature IFfiltering and amplification stage connected to the second mixer.
 3. Aconcurrent, dual-band, image-rejection downconverter for a concurrentdual-band RE receiver having a front-end subsystem that supplies afront-end signal having two discrete desired frequency bands and atransfer function with attentuation regions at frequency bands outsidethe two discrete desired frequency bands, comprising: (a) a firstimage-rejection downconversion stage that receives and that is adaptedto simultaneously downconvert the front-end signal to two intermediatefrequency (IF) bands such that the image frequency bands of the twodesired frequency bands fall at the attenuation regions of the front-endtransfer function; and (b) a subsequent image-rejection downconversionstage that downconverts the two IF bands, further comprising: (i) asecond quadrature local oscillator (LO₂) block adapted produce anin-phase (1) signal and a quadrature (Q) signal at a second givenfrequency; (ii) a first intermediate frequency (IF) mixing stageconnected to the LO₂ block and the first mixer that is adapted to mixthe in-phase intermediate frequency (IF) signal with the in-phase signalof the LO₂ block and to supply a resultant first low frequency (LF)signal; (iii) a second IF mixing stage connected to the LO₂ block andthe second mixer that is adapted to mix the quadrature signal of the LO₂block with the quadrature IF signal and to supply a resultant second LFsignal; (iv) a third quadrature local oscillator (LO₃) block adaptedproduce an in-phase (I) signal and a quadrature (Q) signal at a thirdgiven frequency; (v) a third IF mixing stage connected to the LO₃ blockand the first mixer that is adapted to mix the in-phase LO3 signal withthe in-phase IF signal and to supply a resultant third LF signal; and(vi) a fourth IF mixing stage connected to the LO₃ block and the secondmixer that is adapted to mix the quadrature LO₃ signal with thequadrature IF signal and to supply a resultant fourth LF signal.
 4. Thedownconverter of claim 3, further including a first summing circuit thatcombines the first and second LF signals (to constructively add thefirst desired baseband signal and destructively combine the basebandimage signal associated with the first desired baseband signal); and asecond summing circuit that combines the third and fourth LF signals (toconstructively add the second desired baseband signal and destructivelycombine the baseband image signal associated with the second desiredbaseband signal).
 5. The downconverter of claim 3, wherein the first IFmixing stage includes a first IF mixer, the second IF mixing stageincludes a second IF mixer the third IF mixing stage includes a third IFmixer the fourth IF mixing stage includes a fourth IF mixer.
 6. Thedownconverter of claim 3, wherein the first IF mixing stage includes afirst IF mixer that mixes the in-phase IF signal with in-phase LO₂signal, and a fifth IF mixer that mixes the in-phase IF signal withquadrature LO₂ signal; the second IF mixing stage includes a secondmixer that mixes the quadrature IF signal with the quadrature LO₂ signaland a sixth IF mixer that mixes the quadrature IF signal with thein-phase LO₂ signal; the third IF mixing stage includes a third mixerthat mixes the in-phase IF signal with the in-phase LO₃ signal and aseventh mixer that mixes the in-phase IF signal with the quadrature LO₃signal; and the fourth IF mixing stage includes a fourth mixer thatmixes the quadrature IF signal with the quadrature LO₃ signal and aneighth mixer that mixes the quadrature IF signal with the in-phase LO₃signal.
 7. The downconverter of claim 6, further including a firstsumming circuit that sums the outputs of the first and second mixers; asecond summer that sums tho outputs of the fifth and sixth mixers; athird summing circuit that sums the outputs of the third and fourthmixers; and a fourth summing circuit that sums the outputs of theseventh and eighth mixers.
 8. A concurrent, dual-band, image-rejectiondownconverter for a concurrent dual-band RF receiver having a front-endsubsystem that supplies a front-end signal having two discrete desiredfrequency bands and a transfer function with attentuation regions atfrequency bands outside the two discrete desired frequency bands,comprising: (a) a first image-rejection downconversion stage thatreceives and that is adapted to simultaneously downconvert the front-endsignal to two intermediate frequency (IF) bands such that the imagefrequency bands of the two desired frequency bands fall at theattenuation regions of the front-end transfer function, and furthercomprising: (i) a front-end signal phase shifter connected to thefront-end subsystem that provides a quadrature front-end signal along aquadrature front-end signal path; (ii) a first local oscillator (LO₁)block adapted to supply an phase (I) signal of a first predeterminedfrequency; (iii) a first mixer connected to the LO₁ block and thefront-end subsystem that is adapted to mix the LO₁ signal with thefront-end signal and to supply a resultant in-phase intermediatefrequency (IF) signal; and (iv) a second mixer connected to the LO₁block and the front-end phase-shifter and adapted to mix the LO₁ signalwith the quadrature front-end signal and to supply a resultantquadrature IF signal, wherein the first predetermined frequency of theLO₁ block is offset from the midpoint of the two desired bands such thatthe image frequency bands of the two desired bands fall at attenuationregions of the front-end transfer function; and (b) a subsequentimage-rejection downconversion stage that downconverts the two IF bands.9. A method of concurrently downconverting a dual-band RF signal,comprising: (a) providing a dual-band RF signal having a front-endtransfer function having two desired frequency bands and attenuationnotches at non-selected frequency bands; and (b) downconverting the RFsignal such that the image frequencies of the two-bands fall at theattenuation notches of the front end transfer function, furthercomprising: splitting the RE signal to first and second signalprocessing paths; mixing the RE signal on the first path with anin-phase first local oscillator (LO₁) signal to produce an in-phaseintermediate frequency (IF) signal; filtering the in-phase IF signal;mixing the RF signal on the second path with a quadrature LO₁ signal toproduce a quadrature IF signal; filtering the quadrature IF signal;mixing the filtered in-phase IF signal with an in-phase second localoscillator (LO₂) signal; mixing the filtered quadrature IF signal withthe a quadrature LO₂ signal; mixing the filtered in-phase IF signal withan in-phase LO₃ signal; the filtered quadrature IF signal with thequadrature LO₃ signal; adding the mixed in-phase LO₂ signal to thequadrature LO₂ signal; and subtracting the mixed in-phase LO₃ signalfrom the mixed quadrature LO₃, signal.